Regulated power supply device for a line sweep circuit in a television receiver

ABSTRACT

A regulated power supply device, in particular for a line sweep circuit in a television receiver, containing a chopper circuit (10) which contains in series an inductor (16) and a bidirectional switch (15), which is controlled and mounted in parallel with a capacitor (13), forming a resonant circuit with the inductor (16) when the chopper switch (15) opens. The inductor (16) contains a winding (21), called the power supply winding, of a transformer (20), called the line transformer, in which another winding (22), mounted in series with a reservoir capacitor (33), is connected in parallel with another bidirectional switch (35 and 36), which equips the line sweep output stage (30). The transfer of energy between the chopper circuit (10) and the output stage (30) is done through the windings (21 and 22) of the transformer (20) and its effect is to charge the reservoir capacitor (33) which feeds, apart from the output stage (30), the preceding stages of the sweep circuit (which are not shown). The regulation is done by means of a regulation circuit (40) which receives the line return pulses from an additional winding (25) on the transformer by causing the phase shift to vary between the cut off (opening) instants of the two switches (10 and 35, 36).

The present invention concerns a regulated power supply device, inparticular for a line sweep circuit in a television receiver, which canalso provide D.C. supplies to other circuits in this receiver bysplitting up a D.C. supply voltage which is usually obtained by therectification and filtering of the A.C. mains voltage by means of achopper.

Known chopper converters of this type contain, generally connected inseries between the output terminals of a D.C. power supply source(filtered rectifier), an electronic switch such as a switchingtransistor operating in the saturated and cut off mode and an inductorwhich includes the primary winding of a transformer in which at leastone secondary winding supplies the A.C. energy obtained by the chopping,which is then rectified to provide the D.C. supply voltages with aground insulated from the mains. In most of the known chopper powersupplies, one can vary the output voltages by action on the cyclicratio, i.e. the length of the saturated (closed) state of the switch,for example, by controlling periodically the transistor-chopper by meansof a monostable flip-flop of variable length as a function of a voltagewhich may be picked up at the output of a rectifier fed by a secondarywinding of the transformer so as to form a regulation loop.

Chopper power supplies have frequently been used in television receiversto eliminate the bulky and heavy mains supply transformer and makepossible a regulation of the D.C. power supply voltage for thisreceiver. They have often been combined in particular at the outputstage of the horizontal sweep circuit which supplies them with a pulsesignal at the line frequency that can be used to control the chopping.Various combinations of sweep circuits and chopper power supplies havedescribed, for example, in the French patents or patent applicationswith publication Nos. 2.040.217, 2.060.495, 2.167.549, 2.232.147 or2.269.257, in which the regulation is also done by means of thevariation in the cyclic ratio of the saturated and cut off states of thechopper transistor which, in some cases, is also used as the activeelement of the (final) output stage of the line sweep circuit or of thefeeder stage which controls this circuit.

Chopper power supplies of the so called "pump" type in which the choppertransistor feeds one of the windings of the line transformer during theline return periods and in which the regulation is done by means of thevariation of the internal resistance of this transistor or of a"ballast" transistor in series with this transistor are known, forexample, from the French patents with publication Nos. 2.014.820,2.025.365 or 2.116.335. A circuit of the "pump" type whose choppertransistor has a winding of the line transformer in its collectorcircuit and in which the sweep circuit is electrically insulated fromthe mains has been described in the article by Peruth and Schrenk in theGerman periodical, SIEMENS BAUTEILE REPORT Vol. 12 (1974), No. 4, pages96-98. Its structure corresponds to the contents of the introduction toclaim 1. In circuits of the "pump" type, the chopper transistor or the"ballast" transistor in series with it dissipates an amount of energywhich is not negligable.

In the chopper device supplying power to the output stage of the linesweep circuit with which it is combined in accordance with theinvention, one no longer uses regulation by variation of the internalresistance or of the length of the saturated state of the choppertransistor (or by variation of the cyclic ratio of the chopping with aconstant periodicity) but one does the regulating by variation of therelative phase between the signals of the same frequency which aresupplied respectively by the chopper circuit with a constant cyclicratio and by the output stage of the line sweep, each of which isconnected to one of the windings of a transformer called the linetransformer through which the transfer of energy between the choppercircuit and the sweep output stage takes place as well as in thedirection of the other secondary windings of the line tranformer such asthe very high tension (V.H.T.) winding.

In accordance with the invention, a regulated power supply device, inparticular for a line sweep circuit of a television receiver whichcontains an output stage fitted with a line transformer in which a firstwinding is connected in series with a supply capacitor, is connected inparallel with a first bidirectional switch controlled at the linefrequency, the power supply device containing a chopper circuit with,connected in series between the terminals of a source of a D.C. powersupply voltage, an inductor and a second electronic switch, which canalso be controlled at the line frequency. The inductor in this circuitcontains a second winding of the transformer which is intended for thetransfer of energy between the chopper circuit and the output stage.This power supply device is in particular characterized by the fact thatthe second switch, which is also bidirectional and mounted in parallelwith a tuning capacitor, is so controlled as to be alternately open andclosed during each line period with a constant cyclic ratio and by thefact that the regulation of the power supplied and hence of the voltageat the terminals of the supply capacitor is done by variation of thephase delay between the respective opening instants of the first andsecond switch as a function of the peak amplitude of the line returnpulse for example.

In accordance with a preferred way of making the invention, a powersupply device in accordance with the preceding paragraph, in which thesecond bidirectional switch, which contains a switching transistor, iscontrolled on its base by a regulation circuit in which one input is fedby an auxiliary secondary winding of the line transformer supplying linereturn pulses, is remarkable in particular for the fact that theregulation circuit contains an unstable multivibrator controlling thebase of the chopper transistor and operating independantly on startingup, a circuit generating a variable delay containing a phase shiftstage, which is triggered by the line return pulses and supplies themultivibrator with triggering pulses that are delayed with respect tothe leading edges of the line return pulses, which cause the cut off ofthe chopper transistor, and a regulator stage fed with the line returnpulses and supplying to the phase shift stage a regulation signal whichenables the delay in the triggering pulses to be varied with respect tothe line return pulses as a function of one of the peak amplitudes or ofthe peak to peak amplitude of the line return pulses.

The invention will be better understood and others of itscharacteristics and advantages will appear from the description whichfollows, which is given as an example, and the drawings attached, whichrefer to it. Among them:

FIG. 1 represents part of a theoretical schematic diagram of a chopperpower supply device combined with the output stage of the line sweepcircuit in accordance with the invention;

FIGS. 2a-2f and 3a-3f are diagrams of the voltage wave forms and/orcurrent wave forms at various points in the circuit of FIG. 1 to explainthe operation of this circuit;

FIG. 4 represents part of a synoptic schematic diagram of a simpleproduction model (without a starter device) of regulation circuit 40 inFIG. 1;

FIG. 5 represents a block diagram of a preferred production model ofregulation circuit 40 in FIG. 1 in accordance with the invention;

FIG. 6 represents a theoretical schematic diagram of the whole of thepreferred production model of the regulation circuit in FIG. 5;

FIGS. 7a and 7b represent voltage wave forms illustrating the slaving ofthe frequency of the unstable multivibrator 48 to that of the lineoscillator; and

FIGS. 8a-8c represent voltage wave forms illustrating the operation ofthe regulation by the variation in phase shift.

In FIG. 1 is shown schematically a chopper power supply device of linesweep output stage 30 in accordance with the invention which iselectrically insulated from the A.C. mains which feed rectifier 5 whoseoutput voltage is chopped. This power supply device has two terminals 1,2 which are connected respectively to the two poles of the A.C.distribution mains (220 V, 50 Hz) and feed rectifier diode 3 and filtercapacitor 4, whose capacity is high, which are connected in series andform together a rectifier assembly or a source of D.C. voltage 5. Theoutput of rectifier assembly 5 formed by the two terminals 6 and 7(plates) of the (electro-chemical) capacitor 4 is intended to supply aD.C. power supply voltage V_(A) of the order of 300 V to chopper circuit10. This chopper circuit 10 contains a controlled, bidirectionalelectronic switch 15, which consists of a switching transistor 11 of theNPN type connected with its emitter common and a junction semiconductordiode 12, which are connected in parallel in such a way as to conductrespectively in opposite directions (anti-parallel), and an inductor 16consisting of a choke 14 and a winding 21 of a transformer 20, called aline transformer, connected in series. This winding 21 of linetransformer 20 whose primary winding is normally connected in parallelwith the coils of the horiziontal deviation circuit in the circuit ofline sweep output stage 30 to the supply, through secondary windings,supply voltages in particular to the cathode ray tube will be called inwhat follows the supply voltage winding, because the transfer of energybetween chopper circuit 10 and output stage 30 will be done through it.Switch 15 is mounted in parallel with a capacitor 13 and it is connectedin series with inductor 16 (choke 14 and power supply winding 21 inseries) between the output terminals 6 and 7 of D.C. voltage source 5.This capacitor 13 forms, because of its low capacity with respect tothat of filter capacitor 4, with inductor 16 a parallel, resonant(oscillatory) circuit when electronic switch 15 is opened by the cuttingoff of switching transistor 11 by means of a control signal applied toits base.

Switching transistor 11 is here connected by its collector to one of theterminals of inductor 16, whose other terminal is connected to positiveterminal 6 of source 5 which supplies D.C. power supply voltage V_(A),by its emitter to negative terminal 7 of source 5, which forms a ground,called the primary or hot ground, 8, which is connected to the A.C.mains but is insulated from that 39 of the television set. The base oftransistor 11 is controlled by means of rectangular signals supplied bya regulation circuit 40, which is described further on, in such a way asto be alternately saturated and cut off. Regulation circuit 40 is, forexample, fed by a secondary winding 25 of transformer 20, that suppliessignals whose peak to peak amplitude is proportional to the peakamplitude of the line return pulse. This peak amplitude is a function ofthe energy transfer from chopper circuit 10 to the line sweep outputstage 30 which is connected to another winding 22 of transformer 20.

One may note here that chopper circuit 10 resembles a classical,transistorized, line sweep output stage and that switching transistor 11has been chosen to withstand high collector-emitter voltages (of theorder of 1500 V), and that diode 12 has to withstand the same inversevoltage while switch 15 is open. One may also note that the inductanceof choke 14 may be formed partly or wholly by the leakage inductance ofpower supply winding 21 in transformer 20.

The line sweep output stage 30, which is arranged in classical fashion,contains horizontal deviation coils 31 mounted in parallel and connectedby one of their terminals to a first capacitor 32, called the "forward"or "S effect" capacitor, which feeds them during the forward sweep. Theseries mounting of coils 31 and forward capacitor 32 is connected inparallel, on the one hand, to a second controlled bidirectional switchcontaining a second switching transistor 36 and a second diode 35,called a "shunt" or "parallel" recuperation diode, which are connectedin parallel to conduct in opposite directions, closed (conductor) duringthe forward sweep and open (cut off) during the return sweep, and, onthe other hand, to a second capacitor 34, called the "return" capacitor,which forms, while the second switch is open, a parallel resonantcircuit with the inductance of deviation coils 31. The common point ofthe collector of second transistor 36, of the NPN type, of the cathodeof second diode 35 and return 34 and forward 32 capacitors is connectedto one of the terminals 220 of winding 22 of transformer 20, whichnormally forms the primary winding of this transformer. The otherterminal 221 of winding 22 is connected to one of the terminals of athird capacitor 33 of high capacity, whose other terminal is connectedto the common point of deviation coils 31, return capacitor 34, theanode of second diode 35 and the emitter of second transistor 36, whichis also connected to the ground 39 of the chassis of the televisionreceiver, called the "cold" ground, because it is insulated from theA.C. power supply mains. It is at the terminals of this third capacitor33 that one obtains the D.C. voltage feeding this stage, whose valuedetermines, on the one hand, the peak to peak amplitude of the linesweep current of sawtooth form and, on the other hand, the amplitude ofthe line return voltage pulse which, when rectified after beingtransformed, supplies the very high voltage that polarizes the anode ofthe cathode ray tube (not shown here). The second transistor 36, also aswitching transistor, is controlled by rectangular shaped signalssupplied to input terminals 37 and 38 of stage 30, which arerespectively connected to its base and its emitter, by a feed stage (notshown and called a "driver" in anglo-american literature) so that it isalternately cut off, during the sweep return, and saturated, during thesecond part of the forward sweep.

In classical transistor line sweep circuits, a D.C. voltage sourcegenerally feeds either terminal 221 of winding 22 directly or anintermediate connection to this winding through a diode (see French Pat.Nos. 1.298.087 dated Aug. 11, 1961, 1.316.732 dated Feb. 15, 1962 or1.361.201 dated June 27, 1963) which isolates the primary winding of theline transformer from the D.C. voltage source during the line returninterval.

In the circuit of FIG. 1, it is the A.C. electrical energy transmittedby chopper circuit 10 through windings 21 and 22 of transformer 20 whichcharges capacitor 33 so that it supplies a regulated supply voltage tooutput stage 30. During the line sweep forward periods, when the secondbidirectional switch 35, 36 of sweep output stage 30 is closed(conductor), the terminals of winding 22 of transformer 20 are directlyconnected to those of capacitor 33 which will then receive the energysupplied of by chopper circuit 10.

In FIG. 1, line transformer 20 also has a very high voltage winding 23,one terminal 230 of which may be connected to the ground 39 (or toterminal 220 of winding 22) and whose other terminal 231 is connected tothe input of the very high voltage rectifier assembly or voltagemultiplier (not shown) in classical fashion, and an auxiliary winding 24which may be used to feed either a low voltage rectifier assembly or aload regulator assembly or the filament of the cathode ray tube (notshown). These secondary windings 23, 24 will receive their energy mainlyfrom output stage 30 of the line sweep circuit through winding 22 oftransformer 20, i.e. the line return pulses, the coupling between thewindings will hence be as close as possible.

The operation of the power supply device in FIG. 1 will be explainedbelow with that of output stage 30 of the line sweep circuit, withreference to FIGS. 2 and 3 of the drawing attached, representingdiagrams of the voltage wave forms and/or current wave forms at variouspoints in the schematic diagram of FIG. 1.

In FIGS. 2 and 3, diagram (A) represents the saw tooth wave form of thesweep current i₃₁ (t) in the coils 31 of the horizontal deviationcircuit. Diagram (B) represents the wave form of the voltage v₂₂₀ (t) onterminal 220 of winding 22, which is also that at the terminals of thesecond switch 35, 36. Diagram (C) is the wave form of the voltage v₂₁(t) at the terminals of power supply winding 21 when its leakageinductance is negligable. It is obtained by the transforming of the A.C.component of voltage v₂₂₀ (t). Diagram (D) represents the wave form ofthe voltage v₁₉ (t) at the terminals of first switch 15 in choppercircuit 10, i.e. between the junction 19 of this chopper circuit withinductor 16 and primary ground 8, and diagram (E) represents as a dottedline the current i₁₆ (t) in inductor 16 when output stage 30 is notcontrolled and as a full line the current i₂₁ (t) resulting from thesuperimposition in winding 21 to current i₁₆ (t) on that induced bywinding 22 when output stage 30 is working. Conversely, the current inwinding 22 of transformer 20 results from the superimposition of thecurrent induced by winding 21 on the current produced by the closing ofthe second switch 35, 36, which is analogous to i₃₁ (t) in diagram (A).

The wave forms of diagrams (D) and (E) in FIGS. 2 and 3 are out of phaserespectively, one with respect to another, by a quarter of a line periodT_(H) /4 to allow the illustration of the regulation by the variation inthe relative phase of the voltage v₂₁ and current i₂₁ waves in powersupply winding 21.

The diagrams (F) represent the instantaneous energy E_(i) transmitted bychopper circuit 10 to the output stage 30, which is equal to the productof the wave forms of current i₂₁ (t) and voltage v₂₁ (t) in winding 21,i.e. E_(i) =-v₂₁ ·i₂₁, for two different phase deviations between thevoltage v₂₁ (t) and current i₂₁ (t) waves in power supply winding 21,which correspond respectively to a zero energy transfer in FIG. 2 and amaximum energy transfer in FIG. 3.

The operation of the line sweep output stage 30 is classical once thepower supply capacitor 33 and forward capacitor 32 are charged to a D.C.voltage V₂₂₁ by means of a certain number of chopping cycles, which areindependant on starting up, during which the negative half-cycles of thechopped voltage wave are rectified by recuperation diode 35.

During the forward sweep intervals t_(A), when the switch 35, 36 isclosed from instant t₁ to instant t₃, the current i₃₁ (see A) in thedeviator varies roughly linearly between its negative peak values (att₁) and positive ones (at t₃) with a passage through zero at instant t₂,when current i₃₁ passes from diode 35 to transistor 36, which haspreviously been polarized to conduct. This corresponds to a roughly zerovoltage v₂₂₀ (see B) at the terminals of switch 35, 36.

The line return interval t_(R) is started by the cutting off oftransistor 36 at instant t₃, and the inductance of deviator 31 then actsas a parallel resonant circuit with the return capacitor 34 by causingthe voltage v₂₂₀ (t) to pass through a positive half-sinusoid and reachits peak value at the instant t₄ (or t=0), called the line return pulse,and the current i₃₁ (t) to pass through a half-cosinusoid between thepositive and negative peak values cited, with a passage through zero atthe instant t₄ (or t=0). The mean value of the voltage wave form v₂₂₀(t) at terminal 220 is equal to the D.C. power supply voltage V₂₂₁ atthe terminals of power supply capacitor 33 and forward or S effectcapacitor 32.

The respective peak to peak amplitudes of current i₃₁ (t) (hence thewidth of the screen sweep beam excursion) and of voltage v₂₂₀ (t) (hencethe very high voltage) depend on the value of the D.C. voltage V₂₂₁which feeds the horizontal sweep output stage and which, in most of thechopper power supplies of preceding techniques, is regulated andstabilized by modulating the length of the saturated state (the cyclicratio) of chopper transistor 11 as a function of the amplitude of theline return pulse picked up on an auxiliary winding of line transformer20 (hence of the voltage at the terminals of capacitor 33) and later ofthe rectified and filtered voltage in the network.

In accordance with the invention, the length t_(s) of the saturatedstate of chopper transistor 11 and of the conducting state of diode 12and, as a result, the ratio of this length to that of the complete cycle(line period T_(H)) or to that t_(B) of the cut off state is constantand so chosen as to make the peak amplitude of voltage pulse v₁₉, whichis applied to the collector of transistor 11 during the cut off intervalt_(B), considerably less than its collector-emitter D.C. breakdownvoltage in the cut off state (V_(CEX)) which may exceed 1500 Volts.Thus, for a rectified voltage of 300 V, it is possible to limit thecollector voltage V₁₉ to about 900 Volts by choosing a ratio t_(b)/T_(H) of about 0.5.

As a result, chopper circuit 10 must operate at the line frequency withconduction lengths t_(S) (closed) and cut off lengths t_(B) (open) ofswitch 15 preferably roughly equal (to a line half-period T_(H) /2) andthe regulation of the energy supplied to output stage 30 is done bycausing the respective phases of the line return pulse v₂₂₀ (t) and thecurrent i₂₁ (t) flowing through the power supply winding 21 oftransformer 20 to vary as will be shown further on.

The operation of chopper circuit 10 (fed with D.C. voltage V_(A)) is infact analogous to that of output stage 30, except as far as the formfactor is concerned. This is determined mainly by the respective valuesof the inductance 16 (of choke 14 and the leakage inductance of winding21 of transformer 20 connected in series) and of the capacity of tuningcapacitor 13. The values L₁₆ and C₁₃ are chosen to obtain a half-periodof oscillation slightly less than a line half-period, i.e.: ##EQU1##because the oscillation of the resonant circuit L₁₆, C₁₃ occurs on oneside and on the other of the D.C. voltage V_(A) so that the cut offperiod of chopper switch 15 is greater than this half-period T_(D) /2.

This operation of circuit 10 will first be explained with reference todiagrams D and E in FIG. 2. When, at the instant t=0, transistor 11becomes saturated by a preliminary positive polarization of itsbase-emitter junction, it connects terminal 19 to ground 8 so that acurrent i₁₆ (t) (dotted on diagram E), which is increasing linearly,##EQU2## passes through inductor 16 coming from positive terminal 6 ofpower supply 5.

When transistor 11 receives from regulation circuit 40 a cut off voltageat an instant preceding instant t₆ of the storage time of minoritycharge carries, switch 15 opens and the current stored in inductor 16,i₁₆ (t₆)=V_(A) t₆ /L=V_(A) ·T_(H) /4L, will flow through tuningcapacitor 13 in oscillatory fashion, i.e. cosinusoidally, decreasing toa zero value, while voltage V₁₉ at junction 19 of inductor 16 andcapacitor 13 will increase sinusoidally to a maximum value, these twovalues coinciding in time. Then, capacitor 13 discharges throughinductor 16 also in oscillatory fashion until, at instant t₇, voltagev₁₉ reaches a zero value, which corresponds to a minimum value, i.e.maximum negative, of current i₁₆ (t) whose absolute value is slightlyless than the maximum positive value i₁₆ (t₆). The difference betweenthe absolute peak values i₁₆ (t₆) and i₁₆ (t₇) is explained, on the onehand, by the ohmic losses in circuit 10 and, on the other, by thetransfer of energy between this circuit and, in particular, output stage30.

When oscillatory voltage v₁₉ (t) has exceeded the zero value slightly inthe negative direction, diode 12 starts to conduct so as to connectterminal 19 to ground and produce in inductor 16 a current i₁₆ (t),which increases linearly from its maximum negative value i₁₆ (t₇)towards a zero value where transistor 11, which has already beenpolarized so as to be saturated, picks it up so that it reaches, atinstant t₈, its maximum positive value of instant t₆ again.

It is to be noted here that the mean value of the wave form of voltagev₁₉ at terminal 19 is equal to the D.C. power supply voltage V_(A)between terminals 6 and 7 of filter capacitor 4 in rectifier assembly 5.

If one wishes to obtain an adequate energy transfer between choppercircuit 10 and line sweep output stage 30, it is advantageous to choosethe value of inductor 16 in series with power supply winding 21, i.e.the sum of the leakage inductance of this winding and that of serieschoke 14, so that it is, for example, greater than or equal to threetimes the inductance L₃₁ of the horizontal deviation coils 31, multipledby the square of the transformation ratio between windings 22 and 21,i.e. L₁₆ ≧3l₃₁ (n₁₁ /n₂₁)², and the value of this transformation ration₂₂ /n₂₁ so as to obtain at the terminals of winding 21, during theforward sweep and the closing of switch 15, an induced voltage v₂₁ (t)whose amplitude is between 100 and 150 Volts, i.e. between a third and ahalf the power supply voltage V_(A) at terminals 6, 7 of filtercapacitor 4.

As the D.C. voltage V₂₂₁ at the terminals of capacitor 33 is a functionof the inductance L₃₁ of the horizontal deviation coils 31 and, becauseof this, is between 50 and about 140 Volts, the transformation ratio n₂₂/n₂₁, i.e. between the numbers of turns n₂₂ and n₂₁ of windings 22 and21 respectively, is between 1 and about 4 (preferably between 2 and 3).

The choice of these parameters is only given here as an example, becausethe criterion of this choice is a relative separation between choppercircuit 10 and, in particular, circuit 30 which it feeds, i.e. so thatcurrent i₂₁ (t) in winding 21 is only induced in winding 22 with peakamplitudes which do not exceed about one third those of sweep currenti₃₁ (t) in order not to upset the operation of sweep circuit 30 duringthe conduction of recuperation diode 35. Also, the voltage pulses v₁₉(t) of the diagrams (D) in FIGS. 2 and 3 should not appear at theterminals of winding 21 and should not be transmitted to winding 22 atleast during the opening of sweep switch 36, 35 (line return interval)to winding 22 other than with amplitudes sufficiently small not to upsetthe operation of output stage 30 and the very high voltage rectifier fedby winding 23, while ensuring an energy transfer sufficient to obtain aregulated power supply voltage at the value required.

Transformer 20 may therefore be made in such a way as to have loosercoupling between windings 21 and 22, the self-inductance then consistsof that (L₁₄) of choke coil 14 and the leakage inductance (L₂₁) ofwinding 21. Hence it is advantageous, when one uses a ferrite core(magnetic circuit) of rectangular shape (in the form of a frame), toplace windings 22, 23 and 24 on one of the arms of this core and winding21 and, later, winding 25 on the other. This will also help provide goodinsulation between the primary and secondary grounds 8 and 39. Thedimension of the air gap in the magnetic circuit of transformer 20 or amagnetic shunt, which fixes the leakage inductance L₂₁, and theinductance L₁₄ of the choke 14 are chosen with this result in view.

One may consider then that, from the point of view of the energytransfer from chopper circuit 10 to output stage 30, winding 21 ispassed through by current i₂₁, which consists of triangular shapedcurrent i₁₆ and the current in winding 22, which is induced in saw toothform, superimposed one on the other and that voltage v₂₁, which appearsat its terminals and is shown in diagrams (C) of FIGS. 2 and 3, isroughly analgous to that, v₂₂₀, at the terminals of sweep switch 35, 36but with a mean value of zero.

The energy transmitted by transformer 20 will then be approximatelyequal to the product of voltage v₂₁ (t) and current i₂₁ (t) multipliedby the cosine of the phase angle if one considers the fundamental wavesat the line frequency (15.625 Hz). This is also true for each of theharmonics of the current i₂₁ (t) and voltage v₂₁ (t) waves if onedevelops them in a Fourier series.

The energy ceded duuring each line period T_(H) by chopper circuit 10output stage 30 through transformer 20 may then be written: ##EQU3## Ininductor 16, as a first approximation, current i₂₁ (t) in a sum of anA.C. component i_(A) (t) and a D.C. component I_(c) and, consideringthat the losses of chopper circuit 10 itself are negligable, that themean value of voltage v₂₁ is zero and that the D.C. component I_(c) ofi₂₁ does not take part in the energy transfer, one may write that theenergy supplied by the D.C. source during this period E_(s) =V_(A)·I_(C) ·T_(H) and the A.C. energy supplied by chopper circuit 10,##EQU4## are roughly equal, i.e. ##EQU5## from which it appears thatthere is a mean D.C. current ##EQU6## supplied by source 5 which is aconsequence of the exchange of energy between winding 21 and winding 22in particular. The A.C. energy ceded, E_(H), and, as a result, the D.C.current I_(c) of source 5, varies as a function of the cosine of thephase angle α between each of the respective harmonics of the currenti₂₁ (t) and voltage v₂₁ (t). Hence one can obtain regulation by causingthe phase of the wave of current i₂₁ (t) to vary in power supply winding21 with respect to that of voltage v₂₁ (t) at its terminals to stabilizethe sweep (the peak to peak amplitude of current i₃₁) and/or the veryhigh voltage by acting on the charge supplied to capacitor 33 duringeach cycle.

This is illustrated respectively on the diagrams (F) in FIGS. 2 and 3showing the instantaneous power E_(i) =-v₂₁ (t)·i₂₁ (t) corresponding totwo different phase angles between waves v₂₁ and i₂₁, which indicaterespectively minimum (zero) energy transfers when the zeros of currenti₂₁ coincide with the maxima of voltage v₂₁ or when the respectivemaxima of voltages v₂₁ and v₁₉ are out of phase by a half period T_(H)/2 and maximum energy transfers when the maxima of voltage v₂₁ andcurrent i₂₁ coincide between circuit 10 and output stage 30.

On the diagram (F) in FIG. 2, one can see that, when there is a phasedifference between the corresponding (positive) maxima of v₂₁ (t) andi₂₁ (t) of a quarter of a line period (T_(H) /4) roughly, the energytransfer is zero, because there is equality between the surfaces boundedby the curve and the abscissa, which are respectively above and below itand give a mean value of zero as far as the energy supplied isconcerned.

On the other hand, on the diagram (F) in FIG. 3 in which the product-v₂₁(t)·i₂₁ (t) corresponds to a coincidence of phase between the respectivemaxima of voltage v₂₁ and i₂₁, one can see that, when one subtracts fromthe surfaces above the abscissa the surfaces corresponding to the shadedtriangles below it, three zones remain on the positive side whosesurfaces correspond to the energy which is effectively transferred whosemean value ##EQU7## is positive and shows an effective transfer ofenergy to output stage 30. This translates into a D.C. voltage V₃₃ atthe terminals of capacitor 33 which forms, during the forward sweep(closing of switch 35, 36), the sole load on winding 22 (terminal 220being connected to the ground 39).

Hence, one has shown above that, by causing the phase difference betweenthe corresponding maxima of waves v₂₁ (t) and i₂₁ (t) to vary between 0and T_(H) /4, one can cause the energy transmitted to vary and, as aresult, the voltage V₂₂₁ at the terminals of capacitor 33 which feedsoutput stage 30.

When the relative phase difference between v₂₁ (t) and i₂₁ (t) exceeds aquarter of a line period, as, for example, when the negative peakamplitude of v₂₁ (t) coincides with the negative peak amplitude of i₂₁(t), i.e. a phase difference equal to a line half period (T_(H) /2), theterm of the energy E_(H) becomes negative which indicates that it isoutput stage 30 which feeds chopper circuit 10, or, more precisely,voltage source 5 (capacitor 4). This is not permanently possible unlessit is output stage 30, and hence capacitor 33, which is fed by arectifier assembly, thus showing the reversibility of the power supplydevice in accordance with the invention, which is contrary to classicalchopper power supplies.

Hence, the regulation is done by causing the phase of the opening ofswitch 15 in chopper circuit 10 to be varied by the cutting off oftransistor 11 with respect to the phase of the opening of sweep switch36, 35, which is controlled by the line oscillator (not shown) and isgenerally slaved in frequency and phase to the line synchronizing pulsesof the video complex signal.

Such a variable phase delay is obtained from line return pulses pickedup on one of the windings of transformer 20, such as winding 21 itselfor, as shown in FIG. 1, auxiliary winding 25. These pulses may trigger amonostable flip-flop whose length is variable as a function of the errorvoltage supplied by a comparator in the form of a differentialamplifier, one of whose inputs receives a voltage corresponding eitherto the positive amplitude of v₂₁ (t), which is proportional to thevoltage V₃₃ (V₂₂₁) at the terminals of power supply capacitor 33 inoutput stage 30, or to the peak to peak amplitude of the line returnpulse, which is proportional to the very high voltage, or to acombination of these two criteria. The other input of the differentialamplifier receives a D.C. reference voltage, which may be adjusted, toallow the adjustment of the very high voltage and/or the horizontalsweep current amplitude.

It is to be noted here that power supply winding 21 may be connectedbetween terminal 6 of capacitor 4 and choke 14 in two oppositedirections so that the line return pulses can appear at its junctionwith choke 14 with opposite polarities. Two possibilities of therelative phase of voltage v₂₁ (t) respect to the current i₂₁ (t) inwinding 21 result from this.

In FIG. 4, one has shown a partial block diagram (without a starting updevice) of a simple way of making regulation circuit 40 which controlsthe cut off of transistor 11 in chopper circuit 10 with a delay which isvariable with respect to the line return pulse as a function of thenegative peak amplitude of the signal v₂₅ (t) supplied by auxiliarywinding 25 of transformer 20.

Regulation circuit 40 in FIG. 4 is fed at its first input 401 withsignal v₂₅ (t) supplied by one of the terminals 250 of auxiliary winding25. This signal is roughly the reverse of signal v₂₁ (t) illustrated bythe diagrams (C) respectively in FIGS. 2 and 3 in which onedistinguishes, during each line period, a line return pulse of positivepolarity and a negative plateau whose amplitude is proportional to D.C.voltage V₃₃ at the terminals of capacitor 30. This first input 401feeds, through a first diode 410, the triggering input 411 of a firstmonostable flip-flop 41 of variable length, which produces at its output413, in response to the leading edge of the return pulse, a rectangularsignal whose length varies as a function of the D.C. voltage applied toits length control input 412.

Monostable flip-flops with a pulse length variable as a function of aD.C. voltage are known and a way of making them is described, forexample, in French patent application No. 73.16116 made on May 4, 1973by the present applicant.

This D.C. voltage controlling pulse length is obtained by means of arectifier assembly 42, which is also fed by this first input 401 andcontains a second diode 420 so connected as to conduct only while signalv₂₅ (t) is negative, a capacitor 421 in series with diode 420 whichstores the negative peak values of v₂₅ (t), a resistive potentiometricdivider assembly 422, 423 mounted in parallel with capacitor 421 and apolarity reverser 424 fed by the centre point of divider 422, 423 andsupplying a positive voltage of the same level in reply to a negativeinput voltage, the respective terminals of capacitor 421 and divider422, 423, which are not connected to diode 420, being connected togetherto primary ground 8.

The positive voltage proportional to V₃₃ supplied by reverser 420 feedsa first input 431 receives a stabilized reference voltage, for example,by means of an assembly 44 fed with the mains voltage V₆, rectified andfiltered, through a second input 402 of circuit 40. This assembly 44contains a resistor 440 and a Zener diode 441 connected in seriesbetween the input 402 and primary ground 8 and it supplies, by means ofa resistive divider assembly 442, which may be adjustable and isconnected in parallel with Zener diode 441, the reference voltage toinput 432 of comparator 43. The output 433 of comparator 43, which isconnected to the control input 412 of the first monostable flip-flop 41,supplies it with a voltage proportional to the difference between thevoltages which are applied respectively to its inputs 431 and 432 so asto cause the delay in the cut off of chopper transistor 11 to vary withrespect to that of sweep transistor 36 (FIG. 1) in order to stabilizethe D.C. power supply voltage V₃₃ of output stage 30.

The leading edges of the pulses supplied by output 413 of flip-flop 41coincide roughly with those of the line return pulses and their rear orfalling edges, which occur with variable delays with respect to theleading edges, are used to trigger, eventually through an inverter stage450, a second monostable flip-flop 45 whose output feeds the base ofchopper transistor 11 to cut it off. This second monostable flip-flop 45supplies this base with negative rectangular signals at the linefrequency, of constant length, which is greater than the half period ofoscillation of resonant circuit 13, 15 and hence the half period (>T_(H)/2) and less than three quarters of this same period (<3T_(H) /4) so asto allow transistor 11 to accept the current i₁₆ (t) flowing throughinductor 16 when the current in diode 12 disappears.

FIG. 5 is a block diagram of a preferred production model of aregulation circuit 40 (in FIG. 1) controlling transistor 11 of choppercircuit 10 in accordance with the invention.

In FIG. 5 regulation circuit 40 has an input 401 connected to one of theterminals of auxiliary winding 25 of line transformer 20 which feeds inparallel a first control input 461 of a phase shift stage 46, the inputof a regulator stage 47 and, finally, the input of a synchronizingcircuit 49. The output of regulator stage 47 feeds a second regulationinput 462 of phase shift stage 46, these two stages 46, 47 formingtogether a variable delay generator. The output of phase shift stage 46feeds a first triggering input 481 of an unstable multivibrator 48 whosesecond synchronizing input 482 is fed by the output of synchronizingcircuit 49. This synchronizing circuit 49, whose operation will bedescribed further on, is only necessary if the free running oscillationfrequency of multivibrator 48 is greater than the line frequency. Ifthis is not so, multivibrator 48 is synchronized in classical fashion bythe triggering pulses applied to its input 481. The output of unstablemultivibrator 48 feeds the input of a driver or control stage 50 formedby an amplifier. The output of control stage 50 (called a "driver" inanglo-american litterature), which is connected to output 402 ofregulation circuit 40, feeds the base of transistor 11 in choppercircuit 10.

Auxiliary winding 25 supplies to input 401 of the regulation circuit avoltage wave form containing the line return pulses with a negativepolarity, for example, similar to that shown in the diagrams (C) ofFIGS. 2 and 3. These line return pulses, when applied to input 461 ofphase shift stage 46 or the delay generator, control the triggering of asignal generator which supplies a voltage in the form of a positive sawtooth that is applied to one of the inputs of a voltage comparator stagewhose other input is fed with a fixed reference voltage and whichswitches from its "high" state to its "low" state when the amplitude ofthe saw tooth voltage exceeds the value of the reference voltage.Regulation stage 47 also receives the line return pulses, rectifies themand transmits to regulation input 462 of phase shift stage 46 a signalin the form of a current which enables the slope of the saw tooth to bemodified as a function of the amplitude of the line return pulse whichis a function of the D.C. voltage at the terminals of power supplycapacitor 33 (FIG. 1) in output stage 30. To obtain regulation ofvoltage V₃₃, the phase shift must increase with the value of thisvoltage to regulate the transfer of energy between circuits 10 and 30.As a result, the slope of the saw tooth must decrease with the increasein amplitude of the return pulse. The comparator stage of phase shiftcircuit 46 feeds triggering input 481 of unstable multivibrator 48 totrigger it with a variable phase shift with respect to the leading edgeof the return pulse, which corresponds to the energy transfer desired.Unstable multivibrator 48 is, preferably, synchronized in frequency withline sweep output stage 30 in a way which will be explained later bymeans of synchronizing circuit 49 which feeds its synchronizing input482. The output of multivibrator 48 feeds the input of driver stage 50for chopper transistor 11.

To enable the chopper circuit 10 to start up before the line sweepcircuit is running and, in particular, its output stage 30, unstablemultivibrator 48 must oscillate independantly and stage 50 must amplifythe roughly square wave signal it supplies. For this purpose, anindependant D.C. power supply voltage source 51 is connected to supplyterminals 1, 2 of the A.C. mains and the voltage it supplies feed supplyterminals 403, 404 and 405 of regulation circuit 40. When choppercircuit 10 starts operating independantly when the line sweep circuitcontaining in series a line oscillator, a driver stage and output stage30 is not being fed, the chopper current i₁₆ (t) passing through powersupply winding 21 is induced in winding 22 and it is rectified by thesecond diode 35 which charges positively power supply capacitor 33 whichthen also feeds the other stages of the sweep circuit with a D.C.voltage so that they start up. This starting up and the resultingregulation will be explained more in detail in what follows.

FIG. 6 is a theoretical schematic diagram of the preferred productionmodel of regulation circuit 40 whose block diagram was shown in FIG. 5.

In FIG. 6, power supply voltage source 51 of regulation circuit 40contains a rectifier assembly 52 of the voltage doubler type operatingon a half wave with two diodes 521, 522 in series. The first diode 521is connected by its anode to the second terminal 2 of the supply fromthe mains, which is connected to the primary ground 8 and by its cathodeto the anode of the second diode 522 whose cathode is connected to thepositive plate of a first chemical filter capacitor 523. The negativeplate of the first filter capacitor 523 is connected to the anode of thefirst diode 521 and hence also to primary ground 8. The junction of thecathode of first diode 521 and the anode of second diode 522 is coupledto the first terminal 1 of the power supply from the mains through acoupling capacitor 520 which transmits to the rectifier assembly 52 themains voltage and whose capacity is chosen as a function of the D.C.voltage desired (the voltage drop at the terminals of this capacity 520of the order of a few microfarads makes it possible to obtain arectified and filtered voltage of about 15 Volts). The junction of thepositive plate of first filter capacitor 523 is connected to thepositive plate of a second filter capacitor 524 through a resistor 525,the negative plate of this second capacitor 524 being connected toprimary ground 8. The positive terminal of this second capacitor 524supplies a first rectified and filtered voltage V_(F), on the one hand,through the first output terminal 510 of source 51 to the first positivepower supply terminal 404 of regulation circuit 40 and, on the otherhand, to a stabilizing assembly 53 containing in series a resistor 531and a Zener diode 530 whose anode is connected to primary ground 8. Thejunction of resistor 530 with the cathode of Zener diode 530 isconnected to the second output 511 of source 51, which supplies a secondregulated voltage V_(R) that feeds the second power supply input 403 ofregulation circuit 40.

The first power supply input 404, which supplies a first voltage V_(F)(15 V) that is higher than the second regulated voltage V_(R) (5 V),only feeds control stage 50 of chopper transistor 11. Control stage 50contains in series a phase shift stage 500 (called a "phase splitter" inanglo-american litterature) and an output stage 550 of the "seriespush-pull" type often used in integrated logic circuits of the TTL type.Phase splitter 500 contains a first NPN transistor 501 whose collectoris connected through a collector resistor 502 to the first power supplyinput 404 and whose emitter is connected through an emitter resistor 503to primary ground 8 through the third power supply terminal 405 ofcircuit 40. The base of transistor 501 is connected to the output ofunstable multivibrator 48 through a diode 504 and to the second powersupply input 403 through a polarizing resistor 505. Output stage 550contains a second and third NPN transistors 551 and 552. The collectorof the second transistor 551 is connected through a resistor 553 to thefirst power supply input 404, its base being connected to the collectorof the first transistor 501. The emitter of the second transistor 551 isconnected to the anode of a diode 554 whose cathode is connected to thecollector of the third transistor 552. The base of the third transistor552 is connected to the emitter of the first 501 and its emitter,through the third power supply terminal 405, to primary ground 8. Thejunction of the cathode of diode 554 with the collector of thirdtransistor 552 is connected to the cathode of a Zener diode 555 and tothe positive plate of a chemical capacitor 556, mounted in parallel toform a "battery" which facilitates the cutting off of switchingtransistor 11. The other terminal of the parallel assembly 555, 556 isconnected, through an inductor 557 (choke) to the output 402 ofregulation circuit 40, which feeds the base of switching transistor 11.

Control stage 50 is controlled by an unstable multivibrator 48 of thesymmetrical type containing two NPN transistors 480, 483 mounted withtheir emitters common, i.e. with their emitters connected through thethird power supply terminal 405 to primary ground 8. The collectors ofthe two transistors 480, 483 are connected respectively to the secondpower supply input 403, which receives the stabilized voltage V_(R),through two collector resistors 484, 485. The bases of the twotransistors 480, 483 are connected respectively by means of twopolarizing resistors 486, 487 also to the second power supply input 403.The base of first transistor 480 is also coupled to the collector ofsecond transistor 483 through a first capacitor 488 and the base ofsecond transistor 483 is coupled to the collector of the first 480through a second capacitor 489. The respective values of the polarizingresistors 486, 487 and of the mutual coupling capacitors 488, 489(crossed) of the two stages mounted with their emitters commondetermine, with the value of the stabilized power supply voltage V_(R),the lengths of the half periods of relaxation of multivibrator 48 whosesum (60 μsec) is chosen, preferably, less than that of a line period (64μsec).

In the absence of line return pulses coming from the line sweep outputstage 30 through auxiliary winding 25, multivibrator 48 is fed neitherat its triggering input 481, which is connected to the cathode of afirst diode 4802 whose anode is connected to the base of the secondtransistor 483, nor at its synchronizing input 482 which is connected tothe cathode of a second diode 4803 whose anode is connected to the baseof the first transistor 480. It will operate independantly then as soonas voltage is applied to the mains power supply terminals 1, 2 whichfeed, on the one hand, rectifier assembly 5 and, on the other,independant power supply 51. The power supply then providesmultivibrator 48 with a stabilized power supply voltage V_(R) and thedriver stage 50 with a rectified filtered voltage V_(F). Whenmultivibrator 48 starts to oscillate, it supplies at its output formedby the collector of its second transistor 483 rectangular signals of twolevels (V_(R) and V_(CEsat)), the lowest of which, through couplingdiode 504, causes the cut off of the first transistor 501 in controlstage 50. When the first transistor 501 is cut off, the base of thesecond transistor 551 in output stage 50 is connected, through thecollector resistor 502, to the first power supply input 404 in circuit40 so as to saturate it. The emitter current of second transistor 551then passes, through the diode 554, the Zener diode 555 and inductor 557(which limits the rate of rise of the current di/dt), in resistor 19connecting the base of chopper transistor 11 to primary ground 8 and inthis base in order to allow the saturation of chopper transistor 11, thethird transistor 552 then being cut off by the cut off of the first 501.The voltage drop at the terminals of Zener diode 555 enables thepositive polarizing voltage of the base to be reduced and the capacitor556 to be charged to the Zener voltage V_(Z) during its periods ofconduction.

When the second transistor 483 of multivibrator 48 has switched from itssaturated to its cut off state, its collector voltage is equal to thestabilized voltage V_(R) and diode 504 cuts off. The base of firsttransistor 501 in control stage 50 is then connected to the second powersupply input 403 (+V_(R)) through resistor 505, which causes it tosaturate. Then the emitter current of this first transistor 501 feedsthe base of the third transistor 552 which also becomes saturated whilethe second transistor 551, whose base is at a voltage (V_(CEsat) 501+V_(BE) 552), which is roughly equal to that of its emitter (V_(F) 554+V_(CEsat) 552), cuts off. The saturation of the third transistor 552first brings the base of chopper transistor 11 to a negative voltagewith respect to its emitter V_(BE) 11 =-V_(Z) +V_(CEsat) 552 so as tocut it off rapidly by a rapid evacuation of the minority carriers in itsbase, this voltage V_(BE) 11 then tending asymptotically to zero becausethe capacitor 556 discharges through resistor 19 and the thirdtransistor 552 saturated. Chopper transistor 11 will remain cut offduring the whole half period of oscillation of the resonant circuit L₁₆,C₁₃ and will only accept the current of diode 12 afterwards if it isalready positively polarized on its base by the switching ofmultivibrator 48 to the state in which its second transistor 483 againbecomes saturated so as to cut off first transistor 501 and againsaturate second transistor 551 in control circuit 50.

The alternate cut off and conduction of bidirectional switch 15 causesthe appearance at terminal 19 of recurrent half sinusoids of voltage,shown by the diagrams (D) in FIGS. 2 and 3, a fraction of which is alsopresent at the terminals of power supply winding 21 of line transformer20, from where they are transmitted with a phase inversion (polarity)but without a D.C. component to winding 22 of line sweep output stage30. The negative half cycles of its wave forms on terminal 220 of thewinding are then rectified by the parallel ("shunt") recovery diode 35whose current charges power supply capacitor 33 until the voltage V₃₃ onterminal 221, which feeds the whole of the line sweep circuit, issufficient for the line oscillator (which is not shown) to startoscillating independantly, so as to control, through the driver stage(not shown), switching transistor 36 in output stage 30. Line sweepoutput stage 30 then starts to supply, at the terminals of winding 22 ofline transformer 20, line return pulses v₂₂₀ (t), which are illustratedby the diagrams (B) in FIGS. 2 and 3. These pulses are transmitted toauxiliary winding 25 without a D.C. component and with (negative) phaseinversion so as to have a wave shape analogous to that of the diagrams(C) in FIGS. 2 and 3, which makes possible first the synchronization ofmultivibrator 48 with the line oscillator frequency using an originalslaving device which will be described further on and then theregulation of voltage V₃₃ by varying the delay between the leading edgesof the line return pulses and the instant when chopper transistor 11 inswitch 15 is cut off.

When multivibrator 48 and the line oscillator operate independantly andat different frequencies, this produces a beat because there are randomphase variations between the line return pulses, v₂₂₀ (t) or v₂₁ (t),and the wave form of the chopper voltage v₁₉ (t), so that the energysupplied (or consumed) by chopper circuit 10 to (or from) output stage30 varies from one cycle to another. This has as visible result a moreor less big fluctuation in the amplitude of the line return pulses v₂₂₀(t) which seem to be modulated in amplitude by a sinusoidal signal whosefrequency is equal to the difference between that of multivibrator 48and that of the line oscillator.

If one chooses to synchronize unstable multivibrator 48 in classicalfashion soleby by means of periodic control pulses derived from the linereturn pulses through a variable delay circuit allowing regulation, itis sufficient for the independant oscillation frequency to be less thanthat of the line oscillator. One then obtains on starting up peakvoltages V₁₉, which are higher (overvoltages) on the collector oftransistor 11 when it is cut off because, in the formula V_(19max)·t_(B) =V_(Amax) ·T_(48A), in which V_(19max) is the peak amplitude ofthe collector voltage (on terminal 19), t_(B) the time during whichswitch 15 is cut off, V_(Amax) the maximum supply voltage supplied byrectifier 5 and T_(48A) the free running period of multivibrator 48,T_(48A) being greater than T_(H). If one accepts this overvoltageV_(19max) and limits it by a choice of the saturation time t_(S)slightly higher than the cut off time t_(B1) which is always equal tothe half period of oscillation of L₁₆ and C₁₃, it will not be necessaryto slave multivibrator 48 before regulation and synchronizing circuit 49can be omitted.

If, on the other hand, one wishes to avoid the excesses of the collectorpeak voltage V_(19max) on starting up, one chooses a free running periodT_(48A) for multivibrator 48 less than the line period T_(H) (64 μsec)and one synchronizes by acting only on the length of the cut off stateof first transistor 480 in multivibrator 48 by lengthening it. Duringthis same time interval, second transistor 483 of multivibrator 48 andsecond transistor 551 of driver stage 50 are saturated and the first 501and third 552 transistors of this stage 50 are cut off so that the baseof chopper transistor 11 is polarized to conduct.

This lengthening is done by means of a network 49 containing a diode 490whose cathode is connected to the input 401 of regulation circuit 40which receives the line return pulses from winding 25 with negativepolarity and no D.C. component. The anode of diode 490 is connected tothat of a Zener diode 491 whose cathode is connected to one of theterminals of a first resistor 492. The other terminal of this firstresistor 492 is connected, on the one hand through a second resistor493, to the synchronizing input 482 of unstable multivibrator 48 and, onthe other hand through a third resistor 494, to the collector of thesecond transistor 483 in the multivibrator so that the line returnpulse, negative and with its base cut off by Zener diode 491, cannot acton the base of the first transistor 480 during its periods of saturationso as to cut it off at the wrong time.

The process of slaving the frequency of multivibrator 48 by means of theline return pulses is shown by the diagrams of the wave forms in FIG. 7.

In FIG. 7, the diagram A represents the wave form at the terminals ofauxiliary winding 25 of the line transformer 20 where line return pulsesappear in the form of negative half sinusoids of amplitude V₂₅ at theline frequency (15.626 Hz). The diagram B shows the wave form of thevoltage v_(BE) 480 on the base of the first transistor 480. This waveform contains a first time interval t_(SA) during which chopper switch15 is conducting and transistor 480 is cut off. This time intervaldepends solely on the value of the components connected to this base,specifically the resistor 486 and the capacitor 488 and the supplyvoltage V_(R) for this resistor 484. This wave form also contains asecond time interval t_(B) of fixed length during which chopper switch15 is cut off and transistor 480 saturated. The sum of the intervalst_(SA) and t_(B) represents the period of independent operation T_(A) ofmultivibrator 48 (of the order of 58 μsec for example).

In FIG. 7 the first three periods of free running operation ofmultivibrator 48 are not changed because either the line return pulseoccurs outside the cut off interval t_(SA) of transistor 480 or itsamplitude, with its base cut off by Zener diode 491 and reduced by theresistive voltage divider 492, 494, i.e. (V₂₅ -V_(Z) 491)·R₄₉₄ /(R₄₉₂+R₄₉₄), is less in absolute value than the instantaneous base-emittervoltage v_(BE) 480 (t). From the instant at which the cathode of theseparator diode 4803 becomes more negative than its anode, which isconnected to the base of transistor 480, it begins to conduct a currentI₄₉₃ which discharges capacitor 488 through the resistor 493 in serieswith the resistors 492 and 494 in parallel. Current I₄₉₃ must besubtracted from the current I₄₈₆, which is charging the capacitor,during the whole of the time the amplitude of the line return pulseexceeds the voltage v_(BE). The effect of this is to shift in time apart of the charging wave form of capacitor 496 and thus lengthen thecut off time t_(SA) of transistor 480 by a time Δt_(S) which willincrease until the lengthened period of multivibrator 48 is equal to theline period T_(H). Because the conduction time of switch 15 islengthened, the energy stored in inductor 16 increases. This increasesthe voltage V₃₃ and the amplitude of the line return pulse.

The process of slaving multivibrator 48 in frequency must of necessitylead to equality of these periods because an inequality gives rise to avariation in the peak amplitude of the line return pulse in a directionwhich affects the length of cut off time t_(SA) +Δt_(S) of transistor480 in the opposite direction.

After the slaving of the frequency of unstable multivibrator 48 one cango on to the regulation by varying the phase shift between therespective cut off instants of the sweep transistor 36 and choppertransistor 11 by means of the phase shift 46 and regulator 47 stages inregulation circuit 40, which together form the variable delay generator.

Phase shift stage 46 contains a saw tooth generator which includes afirst capacitor 460, one of whose terminals is connected to primaryground 8 while the other terminal is connected to one of the terminalsof a first resistor 463 whose other terminal is connected to the secondpower supply input 403 which receives the stabilized voltage +V_(R), anda switch, which is intended to short-circuit the first capacitor 460periodically. This switch contains a first NPN switching transistor 464whose collector is connected to the junction of first capacitor 460 andfirst resistor 463, its emitter being connected to primary ground 8 andits base, through a second resistor 465, to the second power supplyinput 403 and, through a third resistor 466, to the anode of a diode467, whose cathode is connected to the control input 461 of phase shiftstage 46 which receives negative line return pulses from input 401 ofcircuit 40. The base of first transistor 464 is also coupled to primaryground 8 through a second capacitor 468.

When input 401 of circuit 40 receives a negative line return pulse,diode 467 starts to conduct and its current causes voltage drops at theterminals of resistors 465, 466 in series which brings transistor 464 tocut off by polarizing it negatively. Second capacitor 468 then chargesto a negative voltage which will extend the length of the cut off oftransistor 464 beyond the disappearance of the line return pulse for apart of the forward sweep period in order to have a sufficientregulation range available.

When the negative return pulse ceases, diode 467 cuts off and secondcapacitor 468 is charged gradually through resistor 465 to a positivevoltage V_(BE) of about 0.7 Volts, at which transistor 464 becomessaturated and discharges first capacitor 460.

During the cut off period of first transistor 464, first capacitor 460is charged almost linearly through resistor 463 and supplies a voltageof positive saw tooth shape to the base of a second NPN transistor 469,whose collector is connected, through a fourth resistor 4600, to thesecond power supply terminal 403 (V_(R) =+5 V). The emitter of secondtransistor 469 is connected, on the one hand, to the cathode of a Zenerdiode 4601 whose anode is connected to primary ground 8 and, on theother hand, to the second power supply terminal 403 through a fifthresistor 4602 which makes it possible to polarize the emitter of secondtransistor 469 at a fixed voltage V_(Z) (between 2 and about 3 Volts).

Second transistor 469 forms, with resistors 4600, 4602 and Zener diode4601, an analog voltage comparator stage which is cut off until thevoltage applied at its base exceeds a threshold voltage resulting fromthe addition of Zener voltage V_(Z) of diode 4601 to the voltage V_(BEm)of about 0.7 Volts at which second transistor 469 saturated.

When second transistor 469 passes from its cut off state to itssaturated state, its collector voltage v_(C) 469 changes from V_(R) toV_(Z) +V_(CEsat). This negative change is transmitted through a couplingcapacitor 4603 to the triggering input 481 of unstable multivibrator 48which is connected, on the one hand, to the cathode of the first diode4802 whose anode is connected to the base of the second transistor 483and, on the other hand, to the first terminals of two resistors 4800 and4801 which form a resistive voltage divider and whose second terminalsare respectively connected to primary ground 8 and to the second powersupply terminal 403 of circuit 40. This negative change, whentransmitted to the base of second transistor 483 in multivibrator 48,causes it to cut off and, in the manner already described, the coppiceof chopper transistor 11 also.

The regulation of the power transmitted by chopper circuit 10 to linesweep output stage 30 is obtained by the variation of the phase shiftbetween the respective cut off instants of the sweep 36 and chopper 11transistors by means of the regulator stage 47 which causes the chargingvoltage slope of the capacitor 460 to vary as a function of one of theparameters contained in the line return pulse.

The combined operation of the phase shift 46 and regulator 47 stageswill be explained by means of FIG. 8, which illustrates the voltage waveforms at three points of these circuits 46, 47.

Regulator stage 47 contains a diode 470 whose cathode is connected tothe input 401 of circuit 40, which receives the negative polarity linereturn pulses and whose anode is connected to the negative plate of afilter capacitor 471 and to one of the terminals of a resistive voltagedivider containing a potentiometer 472 between two resistors 473, 474 inseries and to the anode of a Zener diode 475. The cathode of Zener diode475 is connected, on the one hand, to one of the terminals of a thirdresistor 477 whose other terminal is connected to primary ground 8 and,on the other hand, to the emitter of an NPN transistor 476 whose base isconnected to the slider arm of potentiometer 472 and whose collector isconnected to the regulation input 462 of the phase shift stage 46, whichis connected to the junction of its first capacitor 460 with its firstresistor 463 and the collector of its first transistor 464.

Diode 470 forms with capacitor 471 a rectifier of the negative peaks ofthe line return pulses, capacitor 471 supplying at its terminals avoltage which is a function of the negative peak amplitude of the linereturn.

This rectified peak voltage is applied, on the one hand, to theresistive divider assembly, 472-474, so that the slider arm ofpotentiometer 472 supplies a voltage which is a predetermined adjustablefraction of that voltage and, on the other hand, to the series assemblyof Zener diode 475 and resistor 477 which polarizes this diode 475. Assoon as the amplitude of the line return pulses exceeds the Zenervoltage V_(Z) of diode 475, it is opened up so as to supply at itscathode a voltage equal to the difference between the rectified peakvoltage and the Zener voltage V_(Z). The cathode voltage of Zener diode475 polarizes the emitter of transistor 476 whose base is polarized bydivider assembly 472-474 and which starts to conduct as soon as thefraction of the rectified voltage supplied by the slider arm of thepotentiometer is greater than the Zener voltage V_(Z) in absolute value.Transistor 476 then forms a source of constant current proportional toits base-emitter voltage V_(BE), i.e. to V_(B) -V_(Z) when the latter ispositive. The collector current of transistor 476 is therefore a currentwhich discharges capacitor 460 during the intervals when transistor 464is cut off so as to reduce the slope of the saw tooth voltage at theterminals of capacitor 460. The bigger the negative peak voltage of theline return pulses, the more the collector current of transistor 476reduces the slope so as to increase the delay time between the leadingedge of the line return pulse and the instant of change of thecomparator transistor 469 from its cut off to its saturated state.

This is indicated in FIG. 8, in which the diagram (A) shows the voltagewave form v₂₅ (t) at the terminals of auxiliary winding 25 whose linereturn pulses are of three different amplitudes V_(25B), V_(25F) andV_(25N), the diagram (B) represents the voltage wave form at theterminals of capacitor 460 corresponding to these three line returnpulses and the diagram (C) represents the collector voltage v₄₆₉ (t) ofcomparator transistor 469.

In diagram (A) in FIG. 8, the first line return pulse is of a relativelysmall amplitude V_(25B) which does not cause the conduction ofregulation transistor 476. To this corresponds in diagram (B) thesteepest slope of the voltage wave v₄₆₀ (t) which starts at the instantt₁ of cut off of first transistor 464 in phase shift circuit 46 and theshortest length T_(B) =t₂ -t₁ of this cut off because of the smallernegative charge of capacitor 468. At the instant t₂, when voltage v₄₆₀(t) becomes equal to V_(Z) +V_(BEm), it no longer increases because thediode formed by the base-emitter junction of second transistor 469limits the maximum level of this voltage and transistor 469 becomessaturated. This is illustrated by the diagram (C) in FIG. 8, in whichone can see that the collector voltage v_(C) 469 of second transistor469 contains a negative square wave whose level is equal to V_(Z)+V_(CEsat) and which lasts until the instant t₃ of the opening up of thefirst transistor 464 which discharges capacitor 460 and, as a result,cuts off second transistor 469.

Because of the small phase delay t_(RB) =t₂ -t₁ produced by the fastrise of the voltage v₄₆₀ (t), chopper circuit 10 supplies maximum energyto output stage 30 in the form of a high voltage V₃₃ at the terminals ofthe power supply capacitor 33. As a result, the next line return pulsewill be of large amplitude V_(25F). The comparator transistor 476 startsto conduct as soon as V_(BE) becomes positive and the greater theamplitude V_(25F) to which the capacitor 471 charges, the greater thecollector current. This collector current is to be subtracted from thecharging current of capacitor 460 through the resistor 463. Hence, itcauses a noticeable reduction in the slope of the rise in the voltagev₄₆₀ (t) which occurs between the instants t₄ and t₅. The length of thisrise, which corresponds to the phase delay t_(RF) =t₅ -t₄, will then benoticeably longer than before as well as the length of the cut off stateT_(F) of the first transistor 464. One can see then in the threediagrams that, when V_(25F) is large, the delay t_(RF) is longer and thelength of the negative pulse T_(F) -t_(RF) is slightly shorter.

This longer delay causes a reduction in the voltage V₃₃ compared withthe preceding cycle in which it was too big and the next line returnpulse (the third) will be of an amplitude V_(25N) greater than V_(25B)and less than V_(25F). It will make it possible to obtain, by means ofthe corresponding collector current of the regulation transistor 476, aslope in which the rise from a voltage V_(CEsat) near zero to a voltageV_(Z) +V_(BEm) is of a length equal to t_(RN) =t₇ -t₆. If the slider armof potentiometer 472 has been so placed that the power supply voltageV₃₃ makes it possible to obtain a very high voltage for the cathode raytube (which is not shown) and/or an amplitude of the horizontal sweepcurrent saw tooth corresponding to their respective nominal values, thenominal amplitude V_(25N) of the line return pulse will be reproducedafterwards in recurrent fashion.

It is to be noted here that one can also use as a regulation criterionthe positive amplitude of the signal v₂₅ (t), i.e. the positive planewhose level is proportional to the power supply voltage V₃₃ by using ananalog phase inverter or another winding of line transformer 20 forexample.

One will note also here that the main advantage of the regulation by thephase shift of a chopper circuit operating with a constant cyclic ratioand frequency, compared with that by the variation of one of them, isformed by the fact that the peak voltage applied to the collector of thechopper transistor, when it is cut off, is a function only of the mainsvoltage.

We claim:
 1. A regulated power supply device, in particular for a linesweep circuit in a television receiver, whose output stage (30) containsa first electronic switch of the bidirectional type (36, 35), controlledperiodically so as to be closed during the forward sweep and open duringthe fly-back, connected in parallel with a first series assemblycontaining line deviation coils (31) and a first capacitor (32), calledthe forward capacitor, which feeds these coils (31) during the closingof the first switch (36, 35), with a second capacitor (34), called thereturn capacitor, which forms a parallel resonant circuit with theinductance in particular of the coils (31) during the opening of thefirst switch (36, 35) and with a second series assembly containing afirst winding (22) of a transformer (20), called the line transformer,and a third capacitor (33), called the power supply capacitor, whichfeeds the first winding (22) with D.C. voltage while the first switch(36, 35) is closed, the power supply device containing a chopper circuit(10) connected between the terminals (6, 7) of a D.C. power supplyvoltage source (5) and containing an inductor, called the chopperinductor, (16) and a second electronic switch (15), which is controlled,mounted in series, this second switch (15) containing a choppertransistor (11) controlled on its base by means of a recurring controlsignal, which is produced by means of the line return pulses picked upon a secondary winding (25) of the line transformer (20), in order to bealternately conducting and cut off during each line period, this chopperinductor (16) containing a second winding (21), called the power supplywinding, of this transformer (20), which is intended for the transfer ofenergy between the chopper circuit (10) and the line sweep output stage(30), and being characterized by the fact that, the second switch (15)being also of the bidirectional type and containing, apart from thechopper transistor (11), which is operating in the saturated and cut offmode, a diode (12) mounted in parallel and in opposition with thistransistor, the chopper circuit (10) contains also a fourth capacitor(13), called the turning capacitor, which forms a resonant circuit withthe chopper inductor (16) during the opening periods of the secondswitch (15) which works with a constant cyclic ratio, the periods beingobtained by means of a control signal which causes the cutting off ofthe chopper transistor (11) and their lengths being constant and greaterthan a half period of resonance of this resonant circuit (13, 16) whoselength may reach about a half of a line period, and by the fact that theregulation of the energy exchanged between the chopper circuit (10) andthe output stage (30) is obtained by the variation of the delay betweenthe respective opening instants of the first (36, 35) and second (15)switches.
 2. A power supply device as in claim 1, characterized by thefact that the transistor (11) in the second switch (15) is controlled bymeans of a regulation circuit (40) fed by an auxiliary winding (25) ofthe transformer (20) which supplies it with a signal one of whose peakamplitudes is proportional to the voltage at the terminals of the powersupply capacitor (33) in the output stage (30), which is recharged bymeans of the chopper circuit (10), and whose peak to peak amplitude isproportional to a very high voltage supplied by another winding (23) oftransformer (20), the regulation circuit (40) causing the delay in theinstant of cut off of transistor (11) to vary with respect to theleading edge of the line return pulse produced by the opening of thefirst switch (36, 35).
 3. A power supply device as in claim 2,characterized by the fact that the regulation by the phase shift betweenthe respective cut off instants is obtained as a function either of thepeak to peak amplitude or of the peak amplitude during the fly back orforward sweep of the signal at the terminals of one of the windings (21or 25) of line transformer (20) by comparing this amplitude to areference voltage and by controlling the delay as a function of thedifference between the voltage corresponding to one of these amplitudesand the reference voltage, in order to stabilize either the sweepamplitude or the power supply voltage obtained by rectifying the linereturn pulse.
 4. A power supply device as in claim 2, characterized bythe fact that the regulation circuit (40) contains an unstablemultivibrator (48) whose output is coupled to the base of choppertransistor (11) by means of a control stage (50) and which operatesindependantly on starting up, a circuit generating a variable delaywhich contains a phase shift stage (46) triggered by the line returnpulses and supplying to the multivibrator (48) triggering pulses whichare delayed with respect to the leading edges of the line return pulses,which cause the cutting off of chopper transistor (11), and a regulatorstage (47), which supplies the phase shift stage (46) with a regulationsignal that makes it possible to vary the delay between the respectiveleading edges of the line return pulses and the triggering pulses as afunction of one of the peak amplitudes or of the peak to peak amplitudeof the signal supplied by the auxiliary winding (25) of the transformer(20).
 5. A power supply device as in claim 4, of the type in which thepower supply capacitor (33) feeds a D.C. voltage to the whole line sweepcircuit, characterized by the fact that the regulation circuit (40) isfed by means of an independant power supply circuit (51) which enablesthe chopper circuit (10) to be started up by the independant operationof the unstable multivibrator (48) in order to start up the power supplyof the line sweep circuit with the chopper voltage induced in the firstwinding (22) of the transformer (20) and rectified by the diode (35)which is part of the first bidirectional switch (36, 35) which chargesthe power supply capacitor (33).
 6. A power supply device as in one ofclaims 4 and 5, characterized by the fact that the phase shift stage(46) contains a delay generator which supplies a voltage, in the shapeof recurrent saw teeth (460, 463) which are triggered by the leadingedges of the line return pulses, to an analog voltage comparator stage(469, 4600, 4601), which supplies at its output negative pulses to thebase of the transistor (483) in multivibrator (48) whose cutting offcontrols the cut off of chopper transistor (11) at instants at which theinstantaneous saw tooth amplitude exceeds a fixed threshold voltage(V_(Z) 4601), and by the fact that the regulator stage (47) contains anassembly (470, 471) rectifying the signal supplied by the auxiliarywinding (25) which feeds a signal generator (476, 475) supplying asignal which modifies, from a predetermined threshold, the saw toothslope as a function of one of the peak amplitudes or peak to peakamplitudes of this signal (v₂₅).
 7. A power supply device as in claim 6,of the type in which the free running operating frequency of theunstable multivibrator (48) is less than the line frequency,characterized by the fact that the unstable multivibrator (48) iscontrolled solely by the negative pulses coming from the comparatorstage (469), which are applied to one (483) of the transistors in themultivibrator (48), whose cut off controls that of chopper transistor(11).
 8. A power supply device as in one of claims 4 to 6, of the typein which the free running operating frequency of the unstablemultivibrator (48) is greater than the line frequency in order to limitthe peak voltage (V_(19max)) on the collector of the chopper transistor(11), characterized by the fact that the transistor (480) in themultivibrator (48), whose state is complementary to that of the choppertransistor (11), is fed on its base through a diode (4803) by asynchronizing stage (49), which supplies negative pulses whose amplitudeis equal to a predetermined fraction of that of the line return pulses,in order to lengthen the cut off state of this transistor (480) untilthe sum of these lengths is equal to the line period.